1. Technical Field of the Invention
The present application is directed to the creation of a secondary radar digital monopulse receiver. The digital concept entails converting complex phasers made up of sum beams and difference beams received from a secondary surveillance radar (SSR) antennae to digital form and performing phase detection and log video detection processes using digital methods.
2. Description of Related Art
One of the principle sensors used to determine the presence and location of civilian aircrafts is SSR. The SSR is a beacon type of radar system employing active retransmission techniques on the aircraft. The ground portion of this system solicits both identification and altitude data from the aircraft. The aircraft responds to the request by transmitting either its identity or altitude data.
It is the ability to obtain three-dimensional location (slant range, azimuth, and altitude) and identification data that has made this radar so useful in air traffic control (ATC) systems. The greater reliance placed on this capability by civil ATC worldwide, has driven radar suppliers and developers to provide a significant number of improvements in the SSR. The most noteworthy, in the context of the present application, is the use of monopulse techniques to improve the azimuth accuracy of the sensor.
Current monopulse equipped SSR use an antennae equipped with types of azimuth patterns such as those depicted in FIG. 1a of the present application. A high gain, directional beam (sum pattern 1 as shown in FIG. 1a) is provided, over which RF energy is radiated to the aircraft. This pattern is also used to receive the aircraft response. The antennae is designed to provide two squinted receive only beams, labeled 3a and 3b in FIG. 1a, one existing on either side of the sum beam 1. These beams are used for reception only and are known as the difference beams.
The ratio of the amplitudes of the aircraft reply signals received in the difference beams 3a and 3b, to that of the sum beam 1, then produces a measure which is related to the angle off boresight (boresight is the null axis of the difference beams in the horizontal plane, wherein element 15 of FIG. 1a is the antennae boresight) of the aircraft transmitting the reply. As shown in FIG. 1 of the present application, the sum beam responses are used for extraction of reply codes transmitted by the aircraft transponder. It should be noted that the use of monopulse techniques have resulted in a dramatic four to one improvement in the accuracy of measurement of aircraft azimuth position.
As shown in FIG. 1, element 17 represents a target angle .phi.. Theoretically, this target angle, also known as the angle off boresight (with boresight being element 15 for example) was calculated utilizing a ratio of the sum and difference amplitude to obtain the angle off boresight. The received signals were summed in a summing device 5; difference signals were generated in a difference device 7; and then by dividing the generated sum .SIGMA. shown by 11 in FIG. 1a, by the difference (.DELTA.) in a dividing device 9, a ratio of the amplitudes of the difference (.DELTA.) and the sum (.SIGMA.) shown by element 13 of FIG. 1a was generated. Then, as shown in FIG. 1b, by plotting the generated ratio (.DELTA./.SIGMA.) 13 versus a target angle .phi., a particular angle off boresight was derived.
In previous systems, the determination of the azimuth angle over the reply was accomplished by computing the ratio of the reply amplitude as measured by the difference (3a and 3b) and sum (1) beams. The determination of the monopulse amplitude ratio (difference/sum), to derive azimuth angle, was implemented by one of two competing methods. The first method used amplitude comparison (a non-coherent technique) to directly produce the desired end result at baseband: EQU Azimuth angle (.phi..sub.m)=k [Log (D)-Log (S)] (1)
wherein D was the difference pattern and S was the sum pattern. This first type of receiver never gained wide acceptance, however, due to the amplitude matching requirements needed for the D and S channels.
The second type of receiver combined the sum and difference signals into two vectors S+jD and S-jD, and then extracted the desired ratio of the difference and sum signals (.DELTA./.SIGMA.) 13 as shown in FIG. 1a, with a phase comparator. The second type of receiver (phase monopulse processing) was the type implemented in the Mode S and Monopulse SSR (MSSR) radars. A conceptual block diagram of such a receiver is shown in FIG. 2a.
The Mode S receiver was a two-channel (alpha and beta) version of the phase monopulse processor discussed previously. The sum and difference amplitudes were combined by hybrid circuit 19 to form a first complex phasor s+jd 20 as shown in FIG. 2a, and through a second hybrid circuit 21, to form a second complex phasor -(d+js) 22 as shown in FIG. 2a. FIG. 2b illustrates a relationship between the two phasors 20 and 22. In the two complex phasors, the phase of the signal contains the relative amplitudes of the sum and difference signals (the sum signal designated by s and the difference signal designated by d).
The complex phasor 20 was then passed to another hybrid circuit 23 to derive the demodulation signal and simultaneously passed to a limiter 25a. Thus, the complex phasor 20, the partial phasor js shown by 24 in FIG. 2a, and the phasor 22 were hard limited by limiters 25a, 25b, and 25c respectively. From limiter 25b, a hybrid circuit 27 was then connected, which produced outputs for input to a phase detector 29a and a phase detector 29b, the phase detector 29a also receiving an input from limiter 25a and the phase detector 29b also receiving an output from limiter 25c as shown in FIG. 2a. Phase detector 29a for the alpha channel, and phase detector 29b for the beta channel, then converted the vectors to baseband. Finally, using a summing device 33 connected to phase detector 29a and phase detector 29b, an output azimuth angle .phi..sub.m 35 was then generated.
These limiters 25a-c ensured that the output of the phase detectors 29a and 29b depended only on the relative phase, and not the amplitude of the two complex phasors. As such, less emphasis was needed for amplitude matching of each phase processing channel (alpha and beta) and the reference signal (-s or js).
FIG. 2a also illustrates two log detectors 31a and 31b, connected to the outputs of the hybrid circuit 23 for the alpha and beta channels respectively. The two log detectors produced Log (S) and Log (D) video (baseband) signals. The Log (S) signal, shown as Log sum video (As) 37 in FIG. 2a, was used to extract the digital data transmitted by the aircraft. The Log (D) signal, shown as Log difference video 39 (Ad) in FIG. 2a, was used for receive sidelobe suppression and beam sharpening purposes. The signal 35 was represented in terms of the Log sum video signal 37 and the Log difference video 39 as follows: ##EQU1##
Using logarithmic amplifiers or detectors such as 31a and 31b, and limiters such as 25a-c in the aforementioned Mode S design were extremely expensive. Further, they were prone to failure and drift, the drift problem being one that required recalibration of the receiver more often than desired. Thus, alternative methods of implementing the Mode S type of receiver, for example, were desired, with goals of reducing costs and improving maintainability of the receiver. Prior to the present application, no acceptable results were obtained.